Sensor threshold circuit

ABSTRACT

There is provided a sensor threshold circuit that makes available a hysteresis width that is not dependent on the change in a threshold point. Since a bias current I B  is generated by a threshold current I T  and a threshold adjusting current I CONT , the threshold point is given by a coefficient A and a coefficient K and the hysteresis width |BH| is given by the coefficient K. Accordingly, when the coefficient K is determined, the hysteresis width |BH| is not dependent on the coefficient A and keeps a constant value. In addition, since the coefficient A depends on the resistance ratio, the threshold point is variable by changing a single resistor. Further, when the coefficient K is determined, the hysteresis width |BH| is also determined to be a single value with no variations in its value, no changes according to temperature, and no changes over time.

TECHNICAL FIELD

The present invention relates to sensor threshold circuits applicable to various types of sensors, and more particularly, to a sensor threshold circuit that determines a threshold point for digitizing a sensor output by use of a product of a sensor output impedance and a bias current.

BACKGROUND ART

FIG. 8 shows a conventional sensor threshold circuit. This sensor threshold circuit includes: a four-terminal sensor 10; a voltage comparator 20; a sensor drive current detecting circuit 30; a sensor bias current generating circuit 50; and a bias current switching circuit 70, and obtains a sensor output voltage V_(SO) (=V_(P)−V_(N)) by use of a sensor input B_(IN) applied from the exterior, for instance, magnetic field. In this configuration, the bias current switching circuit 70 switches between resistors R_(O) and R_(R) to change a sensor bias current I_(B), whereby a digital output voltage V_(O) with a hysteresis characteristic as shown in FIG. 9 is made available in receipt of the sensor input B_(IN) (See Patent Document 1, for example).

FIG. 9 is a circuit diagram indicative of the relationship between the sensor input B_(IN) and the output voltage V_(O) in the sensor threshold circuit with the hysteresis characteristic. When the sensor input B_(IN) is increased, the output voltage V_(O) is decreased from V_(OH) to V_(OL) at a threshold point B_(OP). On the other hand, when the sensor input B_(IN) is decreased, the output voltage V_(O) is increased from V_(OL) to V_(OH) in an opposite manner at a threshold point B_(RP) smaller than threshold point B_(OP). Accordingly, the digital output voltage V_(O) with a hysteresis width |BH| is obtainable.

Next, the operation of the conventional threshold circuit will be described with reference to FIG. 8.

Firstly, the threshold point occurring when a switch SW_(O) is conductive and a switch SW_(R) is opened in the bias current switching circuit 70 of FIG. 8 will be described with reference to FIG. 10. For brief description, FIG. 10 shows a circuit configuration indicative of the four-terminal sensor 10, the voltage comparator 20, and the sensor bias current I_(BO) extracted, with the switch SW_(O) conductive and the SW_(R) opened, from the bias current switching circuit 70 of FIG. 8.

In order to facilitate the analysis, firstly, it is assumed that a resistance value of a resistor R_(S) for detecting a sensor drive current I_(S) is considered to be much smaller than the resistance values of sensor resistors R₁, R₂, R₃, and R₄. It is also assumed that a drive terminal voltage V_(CC)2 be equal to a sensor drive voltage V_(CC), accordingly. The resistance value of a sensor drive current detecting resistor R_(S) may take any value, because the result to be produced later will exhibit that the threshold point is not dependent on the sensor drive voltage V_(CC).

At this point, the current I_(BO) generated by the sensor bias current generating circuit 50 is shown as the following expression (1), where I_(S) represents the sensor drive current. I _(BO) =I _(S) ×R _(S) /R _(O)  (1)

For simplification, a current mirror ratio 1/K_(O) is defined as follows. 1/K _(O) =R _(S) /R _(O)  (2)

Now, as shown in FIG. 10, the following expressions (3a) to (3c) are satisfied, where I₁ represents current flowing through the sensor resistor R₁, I₂ represents current flowing through the sensor resistors R₃ and R₄, V_(P) represents a connection point of the sensor resistors R₁ and R₂, and V_(N) represents the potential of the connection point of the sensor resistors R₃ and R₄. I ₁=(V _(CC) −V _(P))/R ₁  (3a) I ₂ =V _(CC)/(R ₃ +R ₄)  (3b) V _(P) /R ₂ =I ₁+(I ₁ +I ₂)/K _(O)  (3c)

When V_(P) is solved, V _(P) =V _(CC)×[(1+1/K _(O))/R ₁+1/{K _(O)×(R₃ +R ₄)}]/(1/R ₂+(1+1/K _(O))/R ₁)  (4) is satisfied. The voltage comparator 20 switches at the voltage satisfying V_(P)=V_(N), thereby forming the following expression (5). V _(CC)×[(1+1/K _(O))/R ₁1/{K _(O)×(R ₃ +R ₄)}]/{1/R ₂+(1+1/K _(O))/R ₁}=R₄ ×V _(CC)/(R ₃ +R ₄)  (5)

In the four-terminal sensor 10, in response to the sensor input B_(IN) applied from the exterior, it is assumed that the balance be lost among the resistors R₁, R₂, R₃, and R₄, thereby resulting in R₁=R₄=R+ΔR, R₂=R₃=R−ΔR, or R₁=R₄=R−ΔR, R₂=R₃=R+ΔR. The sensor output voltage V_(SO) (=V_(P)−V_(N)) will be generated. Accordingly, if R₁=R₄=R+ΔR and R₂=R₃=R−ΔR are satisfied, the expression (5) is formed as follows: [(1+1/K _(O))/(R+ΔR)+1/{K _(O)×(R−ΔR+R+ΔR)}]/(1/(R−ΔR)+(1+1/KO)/(R+ΔR)=(R+ΔR)/(R−ΔR+R+ΔR)  (6)

Then, ΔR/R satisfying the above expression (6) is solved.

$\begin{matrix} \begin{matrix} {{{\Delta R}/R} = {1/\left\lbrack \left( {2 \times K_{O} \times \left\{ {1 + {1/\left( {2 \times K_{O}} \right)}} \right\}} \right\rbrack \right.}} \\ {\approx {1/\left( {2 \times K_{O}} \right)} \equiv B_{OP}} \end{matrix} & (7) \end{matrix}$

That is to say, ΔR/R satisfying the above expression (7) is the threshold point B_(OP). In this sense, the output voltage from a general sensor ranges from several hundreds microvolts to several tens millivolts, whereas the sensor drive voltage substantially ranges from 1 V to 5 V. Approximation is achieved with K_(O) considered to be a sufficiently great value.

Likewise, the threshold point occurring when the switch SW_(O) is opened and the switch SW_(R) is conductive in the bias current switching circuit 70 of FIG. 8 will be described with reference to FIG. 11. For brief description, FIG. 11 shows a circuit configuration indicative of the four-terminal sensor 10, the voltage comparator 20, and the sensor bias current I_(BR) extracted, with the SW_(O) opened and the switch SW_(R) conductive, from the bias current switching circuit 70 of FIG. 8.

At this point, current I_(BR) generated by the sensor bias current generating circuit 50 is shown as the following expression (8). I _(BR) =I _(S) ×R _(S) /R _(R)  (8)

For simplification, a current mirror ratio 1/K_(R) is defined as follows. 1/K _(R) =R _(S) /R _(R)  (9)

Here, the current is defined as shown in FIG. 11, so the current mirror ratio 1/K_(O), in a case where I_(BO) represents the sensor bias current, is equal to 1/K_(R), and considered in the same manner. ΔR/R is given by the following expression (10), when V_(P)=V_(N) is satisfied. ΔR/R≈1/(2×K _(R))≡B _(RP)  (10)

That is to say, ΔR/R satisfying the above expression (10) is the threshold point B_(RP).

The hysteresis width |BH|, which is produced by switching between the switch SW_(O) and the switch SW_(R) in the bias current switching circuit 70, will now be discussed.

The hysteresis width |BH| is obtained by the following expression (11).

$\begin{matrix} \begin{matrix} {{{BH}} = {{B_{OP} - B_{RP}}}} \\ {= {{{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times K_{R}} \right)}}}} \\ {= {{R_{S} \times {\left( {{1/R_{O}} - {1/R_{R}}} \right)/2}}}} \end{matrix} & (11) \end{matrix}$

FIG. 12 depicts the relationship of the threshold points B_(OP) and B_(RP), the hysteresis width |BH|, and the sensor drive current detecting resistor R_(S) which are obtained from the above expressions (7), (10), and (11).

As shown in FIG. 12, in the conventional sensor threshold circuit, it turns out that when the threshold point is varied by changing the resistor R_(S) of the sensor drive current detecting circuit 30, the hysteresis width |BH| is also varied. This is also exhibited by the above expression (11).

Patent Document 1: JP 2001-108480 A

It is to be noted, however, that when the threshold point is varied by changing the resistor R_(S) as described above in the conventional sensor threshold circuit, the hysteresis width |BH| is also varied in the same manner. The hysteresis width |BH| reduces the variations in the output caused by sensor output noises. Hence, when the threshold point is varied, there is a drawback in that the influences of the output variations caused by the sensor output noises are different depending on the threshold point, due to the variations in the hysteresis width |BH|.

Moreover, when the threshold point is varied by changing the resistor R_(O) and the resistor R_(R), the hysteresis width |BH| that is not dependent on the change in the threshold point is obtainable. However, two output terminals are needed to change the two resistors, thereby leading to another drawback of increasing the chip area and chip costs.

The present invention has been made in view of the above drawbacks, and has an object of providing a sensor threshold circuit that enables variations in a threshold point by changing a single resistor to achieve a hysteresis width that is not dependent on the variations in the threshold point.

DISCLOSURE OF THE INVENTION

According to an aspect of the present invention, there is provided a sensor threshold circuit that outputs a digital signal with a hysteresis characteristic, in receipt of an input from a sensor, the sensor threshold circuit comprising: a voltage comparator that digitizes an output voltage of the sensor; a sensor drive current detecting circuit that detects a sensor drive current; a sensor bias current generating circuit that generates a threshold current of 1/K (where K>0) times of the sensor drive current that has been detected by the sensor drive current detecting circuit, based upon the output from the voltage comparator; a threshold adjusting current generating circuit that generates a threshold adjusting current of 1/A (where A>0) times of the sensor drive current that has been detected by the sensor drive current detecting circuit, adds or subtracts the threshold adjusting current to or from the threshold current that has been generated by the sensor bias current generating circuit to generate a sensor bias current, and supplies the sensor bias current to an output terminal of the sensor.

The above configuration may further include a bias current switching circuit that switches between a first resistor and a second resistor, based upon an output from the voltage comparator, wherein the sensor bias current generating circuit generates the threshold current, based upon the first resistor or the second resistor connected by the bias current switching circuit.

In the above configuration, the threshold adjusting current generating circuit may include: a first resistor for changing a threshold point with a sensor drive voltage used as a reference; second and third resistors for changing the threshold point with GND used as the reference; and an operating amplifier, and said 1/A satisfies a following expression, 1/A=(R_(S)/R_(A)/2)×(1−R_(C)/R_(B)), where R_(A) represents a resistance value of the first resistor, R_(B) and R_(C) (where R_(B)>R_(C)) represent resistance values of the second and third resistors, respectively, and R_(S) represents a resistance value of a resistor in the sensor drive current detecting circuit.

In the above configuration, at least one of the first, second, and third resistors may be a variable resistor.

In the above configuration, the threshold adjusting current generating circuit may include: a first resistor for changing a threshold point with a sensor drive voltage used as a reference; and an operating amplifier, and said 1/A satisfies a following expression, 1/A=R_(S)/R_(A), where R_(A) represents a resistance value of the first resistor, and R_(S) represents a resistance value of a resistor in the sensor drive current detecting circuit.

In the above configuration, the first resistor may be a variable resistor.

In the above configuration, 1/K=R_(S)/R_(O) may be formed when only the first resistor is conductive, and 1/K=R_(S)/R_(R) may be formed when only the second resistor is conductive, where R_(O) represents a resistance value of the first resistor and R_(R) represents a resistance value of the second resistor.

In the above configuration, the sensor may be a four-terminal sensor and may be any one of Hall element, magnetic resistive element, strain sensor, pressure sensor, temperature sensor, and acceleration sensor.

According to an aspect of the present invention, as described above, since a threshold current I_(T) and a threshold adjusting current I_(CONT), which are produced by the sensor drive current, generate a bias current I_(B), the hysteresis width |BH| is given by the resistance ratio K. Accordingly, when a resistance ratio (coefficient) K is determined, the hysteresis width |BH| is not dependent on a coefficient A changing the threshold point, and keeps a constant value. Meanwhile, when the resistance ratio K is determined, the hysteresis width |BH| is also determined to be a single value with no variations in its value, no changes according to temperature, and no changes over time. Therefore, according to an aspect of the present invention, there is an effect of providing a sensor threshold circuit that makes available the hysteresis width |BH| that is not dependent on the variations in a threshold point.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a conceptual view of a configuration of a sensor threshold circuit according to a first embodiment of the present invention;

FIG. 2 illustrates a circuit diagram indicative of a circuit configuration with a four-terminal sensor, a voltage comparator, a threshold current I_(T), and a threshold adjusting current I_(CONT) extracted from FIG. 1;

FIG. 3 illustrates a circuit diagram indicative of a configuration of a sensor threshold circuit according to a second embodiment of the present invention;

FIG. 4 illustrates a circuit diagram indicative of a circuit configuration with a four-terminal sensor, a sensor drive current detecting circuit, and a threshold adjusting current generating circuit extracted from FIG. 3;

FIG. 5 illustrates a circuit diagram indicative of a circuit configuration with the four-terminal sensor, a voltage comparator, a threshold current I_(T), and a threshold adjusting current I_(CONT) extracted from FIG. 3;

FIG. 6 depicts the relationship between a threshold point and a coefficient 1/A according to an embodiment of the present invention;

FIG. 7 illustrates a circuit diagram indicative of a configuration of a sensor threshold circuit according to a third embodiment of the present invention;

FIG. 8 shows a circuit diagram indicative of a conventional sensor threshold circuit;

FIG. 9 shows the relationship between a sensor input and an output voltage of the sensor threshold circuit with a hysteresis characteristic;

FIG. 10 shows a circuit configuration indicative of a four-terminal sensor, a voltage comparator, and a sensor bias current I_(BO) extracted, with a switch S_(WO) conductive and a S_(WR) opened, from a bias current switching circuit of FIG. 8;

FIG. 11 shows a circuit configuration indicative of the four-terminal sensor, the voltage comparator, and the sensor bias current I_(BO) extracted, with the switch S_(WO) opened and the S_(WR) conductive, from the bias current switching circuit of FIG. 8; and

FIG. 12 depicts the relationship of threshold points and a sensor drive current detecting resistor R_(S) in the conventional sensor threshold circuit.

EXPLANATION OF REFERENCES

10 four-terminal sensor, 20 voltage comparator, 30 sensor drive current detecting circuit, 40 sensor drive voltage source, 50 sensor bias current generating circuit, 60 threshold adjusting current generating circuit, 70 bias current switching circuit, 52, 61, 62 operating amplifier, 51, 63, 64 PMOS transistor, 65, 66, 67 NMOS transistor, 71 inverter, SW_(O), SW_(R) switch, R₁, R₂, R₃, R₄, R_(S), R_(O), R_(R), R_(A), R_(B), R_(C), resistor

BEST MODE FOR CARRYING OUT THE INVENTION

The embodiments of the present invention will now be described with reference to the accompanying drawings. In the drawings, like references are used to denote corresponding parts with each other in all of the drawings, and the description for the overlapping parts will be omitted as necessary.

First Embodiment Configuration of First Embodiment

FIG. 1 illustrates a conceptual view of a configuration of a sensor threshold circuit according to a first embodiment of the present invention.

The sensor threshold circuit includes: a four-terminal sensor 10; a voltage comparator 20; a sensor drive current detecting circuit 30; a sensor drive voltage source 40; a sensor bias current generating circuit 50; a threshold adjusting current generating circuit 60; and a bias current switching circuit 70. The four-terminal sensor 10 is a resistance type four-terminal sensor, and is any one of Hall elements, magnetic resistive elements, strain sensors, pressure sensors, temperature sensors, and acceleration sensors, for example.

In the sensor threshold circuit, a sensor bias current I_(B), which has been generated by the sensor bias current generating circuit 50 and the threshold adjusting current generating circuit 60, is supplied to an output terminal of the four-terminal sensor 10, so that a hysteresis characteristic is made available at a sensor output voltage V_(SO) (=V_(P)−V_(N)), by use of a voltage drop resulted from a product of I_(B)×R_(OUT), where I_(B) represents a sensor bias current and R_(OUT) represents an impedance of an output terminal V_(P) of the four-terminal sensor 10.

In the sensor threshold circuit with such a configuration, the sensor drive current I_(S) is detected with the use of the sensor drive current detecting circuit 30, and a threshold current I_(T), which is 1/K (K>0) times of the sensor drive current I_(S), is generated at the sensor bias current generating circuit 50. In this sense, K has two values due to the control of the bias current switching circuit 70, so the sensor bias current generating circuit 50 generates two threshold currents I_(TO) and I_(TR).

The threshold adjusting current I_(CONT), which is 1/A (A>0) times of the sensor drive current I_(S) is produced at the threshold adjusting current generating circuit 60 and the threshold current I_(T) and the threshold adjusting current I_(CONT) are added or subtracted, so as to generate the sensor bias current I_(BO).

A key point of the sensor threshold circuit according to an aspect of the present invention is configured to produce the threshold current I_(T) and the threshold adjusting current I_(CONT) based upon the sensor drive current I_(S), so as to generate the sensor bias current I_(BO) by adding or subtracting the threshold adjusting current I_(CONT) to or from the threshold current I_(T).

In order to facilitate the understanding of the present invention, the threshold current I_(T) and the threshold adjusting current I_(CONT) of FIG. 1 will be discussed firstly.

For simplification of the description, FIG. 2 illustrates a circuit diagram with the four-terminal sensor 10, and the voltage comparator 20, and a sensor drive voltage source 40 extracted from the sensor threshold circuit of FIG. 1, to which the threshold current I_(T) and the threshold adjusting current I_(CONT) are supplied.

A threshold point B_(OP) will be discussed, while the sensor bias current generating circuit 50 of FIG. 1 is being made to generate the threshold current I_(TO) by the bias current switching circuit 70.

At this moment, the sensor bias current I_(BO) is expressed as the following expression (12), where 1/K_(O) represents a current mirror ratio of the sensor bias current generating circuit 50 and 1/A represents a current mirror ratio of the threshold adjusting current generating circuit 60.

$\begin{matrix} \begin{matrix} {I_{BO} = {I_{TO} - I_{CONT}}} \\ {= {{I_{S}/K_{O}} - {I_{S}/A}}} \end{matrix} & (12) \end{matrix}$

Here, the following expressions (13a) to (13c) are formed. I ₁=(V _(CC) −V _(P))/R ₁  (13a) I ₂ =V _(CC)/(R ₃ +R ₄)  (13b) V _(P) /R ₂ =I ₁+((I ₁ +I ₂)/K _(O)−(I ₁ +I ₂)/A))  (13c)

When V_(P) is solved, V_(P) =V _(CC)×[(1+1/K _(O)−1/A)/R ₁+(1/K _(O)−1/A)/(R ₃ +R ₄)]/[1/R ₂+(1+1/K _(O)−1/A)/R ₁]  (14) is satisfied. The voltage comparator 20 switches at the voltage satisfying V_(P)=V_(N), thereby forming the following expression (15). V _(CC)×[(1+1/K _(O)−1/A)/R ₁+(1/K _(O)−1/A)/(R₃ +R ₄)]/[1/R ₂+(1+1/K _(O)−1/A)/R ₁ ]=R ₄ ×V _(CC)/(R ₃ +R ₄)  (15)

In the four-terminal sensor 10, in response to the sensor input B_(IN) applied from the exterior, it is assumed that the balance be lost among the resistors R₁, R₂, R₃, and R₄, thereby resulting in R₁=R₄=R+ΔR, R₂=R₃=R−ΔR, or R₁=R₄=R−ΔR, R₂=R₃=R+ΔR. The sensor output voltage V_(SO) (=V_(P)−V_(N)) will be generated. Accordingly, if R₁=R₄=R+ΔR and R₂=R₃=R−ΔR are satisfied, the expression (16) is formed as follows: {(1+1/K _(O)−1/A)/(R+ΔR)+(1/K _(O)−1/A)/(R−ΔR+R+ΔR)}/{(1/(R−ΔR)+(1+1/K _(O)−1/A)/(R+ΔR)}=(R+ΔR)/(R−ΔR+R+ΔR)  (16)

ΔR/R satisfying the above expression (16) is solved.

$\begin{matrix} \begin{matrix} {{\Delta\; R\text{/}R} = {\left( {{1/K_{O}} - {1/A}} \right)/\left\lbrack {2 \times \left\{ {1 + {{1/2} \times \left( {{1/K_{O}} - {1/A}} \right)}} \right\}} \right\rbrack}} \\ {\approx {{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times A} \right)}} \equiv B_{OP}} \end{matrix} & (17) \end{matrix}$

That is to say, ΔR/R satisfying the above expression (17) is the threshold point B_(OP). In this sense, approximation is performed by utilizing constant numbers K_(O) and A that can be handled as sufficiently great values in the assumed range.

The threshold point B_(RP) will now discussed when the sensor bias current generating circuit 50 of FIG. 1 is made to generate the threshold current I_(TR) by the bias current switching circuit 70.

Here, the sensor bias current I_(BO) is expressed as the following expression (18), where 1/K_(R) represents a current mirror ratio of the sensor bias current generating circuit 50 and 1/A represents a current mirror ratio of the threshold adjusting current generating circuit 60.

$\begin{matrix} \begin{matrix} {I_{BR} = {I_{TR} - I_{CONT}}} \\ {= {{I_{S}/K_{R}} - {I_{S}/A}}} \end{matrix} & (18) \end{matrix}$

The threshold point B_(RP) at this point is obtainable by replacing the current mirror ratio 1/K_(O), of the sensor bias current generating circuit 50, included in the expressions (12) to (17), with 1/K_(R). ΔR/R≈1/(2×K _(R))−1/(2×A)≡B _(RP)  (19)

The hysteresis width |BH|, which is produced when the sensor bias current generating circuit 50 is made to switch between the threshold current I_(TO) and the threshold current I_(TR) by the bias current switching circuit 70, will now be discussed.

The hysteresis width |BH| is expressed by the following expression (20).

$\begin{matrix} \begin{matrix} {{{BH}} = {{B_{OP} - B_{RP}}}} \\ {= {{\left\{ {{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times A} \right)}} \right\} - \left\{ {{1/\left( {2 \times K_{R}} \right)} - {1/\left( {2 \times A} \right)}} \right\}}}} \\ {= {{{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times K_{R}} \right)}}}} \end{matrix} & (20) \end{matrix}$

What is important in the above expressions (17), (19), 20 obtained according to the above embodiments is that the threshold points BOP and BRP are dependent on a coefficient A, but the hysteresis width |BH| is not dependent on the coefficient A.

In addition, the threshold point is variable by changing the coefficient A.

Furthermore, the above expression (20) exhibits that the hysteresis width |BH| is not dependent on the sensor drive voltage V_(CC).

The above coefficients K_(O) and K_(R) are given by the mirror ratio. Accordingly, when the coefficients K_(O) and K_(R) are determined, the hysteresis width |BH| is determined to be a single value with no variations in its value, no changes according to temperature, and no changes over time.

Therefore, the sensor threshold circuit according to an aspect of the present invention achieves the hysteresis width that is not dependent on the change in the threshold point.

It is to be appreciated that the sensor bias current I_(B) produced by the sensor bias current generating circuit 50 and the threshold adjusting current generating circuit 60 is made to be supplied to an output terminal V_(P) of the four-terminal sensor 10 in the first embodiment of the present invention, but may be supplied to an output terminal V_(N) instead of the output terminal V_(P) of the four-terminal sensor 10.

Second Embodiment Configuration of Second Embodiment

FIG. 3 illustrates a circuit diagram indicative of a configuration of a sensor threshold circuit according to a second embodiment of the present invention.

The sensor threshold circuit includes: a four-terminal sensor 10; a voltage comparator 20; a sensor drive current detecting circuit 30; a sensor drive voltage source 40; a sensor bias current generating circuit 50; a threshold adjusting current generating circuit 60; and a bias current switching circuit 70. The four-terminal sensor 10 is a resistance type four-terminal sensor, and is any one of Hall elements, magnetic resistive elements, strain sensors, pressure sensors, temperature sensors, and acceleration sensors, for example.

In the sensor threshold circuit, a sensor bias current I_(B), which has been generated by the sensor bias current generating circuit 50 and the threshold adjusting current generating circuit 60, is supplied to an output terminal of the four-terminal sensor 10, so that a hysteresis characteristic is available at a sensor output voltage V_(SO) (=V_(P)−V_(N)), by use of a voltage drop resulted from a product of I_(B)×R_(OUT), where I_(B) represents a sensor bias current and R_(OUT) represents an impedance of an output terminal V_(P) of the four-terminal sensor 10.

In the sensor threshold circuit with such a configuration, a sensor drive current I_(S) is detected with the use of a resistor R_(S) of the sensor drive current detecting circuit 30, and a threshold current I_(T), which is 1/K (K>0) times of the sensor drive current I_(S), is generated at the sensor bias current generating circuit 50. The sensor bias current generating circuit 50 is composed of: a resistor R_(S); a resistor R_(R) or a resistor R_(O) connected by the control of the bias current switching circuit 70; an operating amplifier 52; and a PMOS transistor 51. In this sense, K has two values due to the control of the bias current switching circuit 70, so the sensor bias current generating circuit 50 generates two threshold currents I_(TO) and I_(TR).

Also, a threshold adjusting current I_(CONT), which is 1/A (A>0) times of the sensor drive current I_(S), is produced at the threshold adjusting current generating circuit 60. The threshold adjusting current generating circuit 60 is composed of: a resistor R_(A); an operating amplifier 61; a PMOS transistor 63; a PMOS transistor 64; a resistor R_(C); a resistor R_(B); an operating amplifier 62; an NMOS transistor 65; an NMOS transistor 66; and an NMOS transistor 67.

The threshold current I_(T) and the threshold adjusting current I_(CONT) are subtracted at the threshold adjusting current generating circuit 60, and then the sensor bias current I_(B) is generated. In the second embodiment, the description will be given of current, in which the threshold current I_(CONT) is subtracted from the threshold current I_(T), is used as the sensor bias current I_(BO). However, the sensor bias current I_(BO) may be current in which the threshold adjusting current I_(CONT) is added to the threshold current I_(T).

A key point of the sensor threshold circuit according to an aspect of the present invention is configured to produce the threshold current I_(T) and the threshold adjusting current I_(CONT) based upon the sensor drive current I_(S), so as to generate the sensor bias current I_(BO) by adding or subtracting the threshold adjusting current I_(CONT) to or from the threshold current I_(T).

In order to facilitate understanding of the present invention, the threshold current I_(T) and the threshold adjusting current I_(CONT) in FIG. 3 will now be discussed. Also, for simplification of the description, it is assumed that the resistance value of the resistor R_(S) for detecting the sensor drive current be considered to be extremely small, as compared to the resistance values of the sensor resistors R₁, R₂, R₃, and R₄, thereby making a sensor drive terminal voltage V_(CC)2 equal to V_(CC). The resistance value of a sensor drive current detecting resistor R_(S) may take any value, because the result to be produced later will exhibit that the threshold point is not dependent on the sensor drive voltage V_(CC).

Firstly, the threshold current I_(TO) occurring when the switch SW_(O) is conductive and the switch SW_(R) is opened in the bias current switching circuit 70 shown in FIG. 3 will be described. I_(TO) =I _(S) ×R _(S) /R _(O)  (21)

For simplification, a current mirror ratio 1/K_(O) of the sensor bias current generating circuit 50 is defined as follows. 1/K=R _(S) /R _(O)  (22)

Next, the threshold current I_(TR) occurring when the switch SW_(R) is conductive and the switch SW_(O) is opened in the bias current switching circuit 70 shown in FIG. 3 will be described. I _(TR) =I _(S) ×R _(S) /R _(R)  (23)

For simplification, the current mirror ratio 1/K_(R) of the sensor bias current generating circuit 50 is defined as follows. 1/K _(R) =R _(S) /R _(R)  (24)

The threshold adjusting current I_(CONT) will be described with the current defined as illustrated in FIG. 4. For simplification of the description, FIG. 4 illustrates a circuit diagram including: the four-terminal sensor 10; the sensor drive current detecting circuit 30; the sensor drive voltage source 40; and the threshold adjusting current generating circuit 60, which are extracted from the sensor threshold circuit of FIG. 3. Here, for brief description, it is assumed that the PMOS transistor 63 and the PMOS transistor 64 have the same electrical characteristics. In addition, it is assumed that the NMOS transistor 66 and the NMOS transistor 67 have the same size electrical characteristics. Further, it is assumed that R_(B)>R_(C) be satisfied.

Currents I₆₃ and I₆₄, which pass across the PMOS transistor 63 and the PMOS transistor 64, respectively, are equal to each other and expressed as the following expression (25). I ₆₃ =I ₆₄ =I _(S) ×R _(S) /R _(A)/2  (25)

Current I₆₅, which passes across the NMOS transistor 65 and a resistor R_(B) is expressed as the following expression (26).

$\begin{matrix} \begin{matrix} {I_{65} = {I_{64} \times {R_{C}/R_{B}}}} \\ {= {I_{S} \times {{R_{S}/R_{A}}/2} \times {R_{C}/R_{B}}}} \end{matrix} & (26) \end{matrix}$

Current I₆₆, which passes across the NMOS transistor 66 and the NMOS transistor 67, and the threshold adjusting current I_(CONT) are equal to each other and expressed as the following expression (27).

$\begin{matrix} \begin{matrix} {I_{66} = I_{CONT}} \\ {= {I_{63} - I_{65}}} \\ {= {\left( {I_{S} \times {{R_{S}/R_{A}}/2}} \right) - {\left( {I_{S} \times {{R_{S}/R_{A}}/2}} \right) \times {R_{C}/R_{B}}}}} \\ {= {\left( {I_{S} \times {{R_{S}/R_{A}}/2}} \right) \times \left( {1 - {R_{C}/R_{B}}} \right)}} \end{matrix} & (27) \end{matrix}$

In this sense, the following expression (28) is defined for simplification. 1/A=R _(S) /R _(A)/2×(1−R _(C) /R _(B))  (28)

Currents are defined as illustrated in FIG. 5. For brief description, FIG. 5 illustrates a circuit diagram including: the four-terminal sensor 10; and the voltage comparator 20, which are extracted from the sensor threshold circuit of FIG. 3, and to which the threshold current I_(T) and the threshold adjusting current I_(CONT) are supplied.

The threshold point B_(OP) occurring when the switch SW_(O) is conducted and the switch SW_(R) is opened in the bias current switching circuit 70 of FIG. 3 will be discussed.

In this sense, the threshold current I_(TO) is expressed as the above expression (21), the sensor bias current I_(BO) is expressed as the following expression (29), by use of the above expressions (21), (22), (27), and (28).

$\begin{matrix} \begin{matrix} {I_{BO} = {I_{TO} - I_{CONT}}} \\ {= {{I_{S}/K_{O}} - {I_{S}/A}}} \end{matrix} & (29) \end{matrix}$

At this point, the following expressions (30a) to (30c) are satisfied. I ₁=(V _(CC) −V _(P))/R ₁  (30a) I ₂ =V _(CC)/(R ₃ +R ₄)  (30b) V _(P) /R ₂ =I ₁+((I ₁ +I ₂)/K _(O)−(I ₁ +I ₂)/A)  (30c)

When V_(P) is solved, V _(P) =V _(CC)×[(1+1/K _(O)−1/A)/R ₁+(1/K_(O)−1/A)/(R ₃ +R ₄)]/(1/R ₂+(1+1/K _(O)−1/A)/R ₁)  (31) is satisfied. The voltage comparator 20 switches at the voltage satisfying V_(P)=V_(N), thereby forming the following expression (32). V _(CC)×[(1+1/K _(O)−1/A)/R ₁+(1/K _(O)−1/A)/(R ₃ +R ₄)]/[1/R ₂+(1+1/K _(O)−1/A)/R ₁ ]=R ₄ ×V _(CC)/(R ₃ +R ₄)  (32)

In the four-terminal sensor 10, in response to the sensor input B_(IN) applied from the exterior, it is assumed that the balance be lost among the resistors R₁, R₂, R₃, and R₄, thereby resulting in R₁=R₄=R+ΔR, R₂=R₃=R−ΔR, or R₁=R₄=R−ΔR, R₂=R₃=R+ΔR. The sensor output voltage V_(SO) (=V_(P)−V_(N)) will be generated. Accordingly, if R₁=R₄=R+ΔR and R₂=R₃=R−ΔR are satisfied, the following expression (33) is formed. {(1+1/K _(O)−1/A)/(R+ΔR)+(1/K _(O)−1/A)/(R−ΔR+R+ΔR)}/{1/(R−ΔR)+(1+1/K _(O)−1/A)/(R+ΔR)}=(R+ΔR)/(R−ΔR+R+ΔR)  (33)

ΔR/R satisfying the above expression (33) is solved.

$\begin{matrix} \begin{matrix} {{\Delta\; R\text{/}R} = {\left( {{1/K_{O}} - {1/A}} \right)/\left\lbrack {2 \times \left\{ {1 + {{1/2} \times \left( {{1/K_{O}} - {1/A}} \right)}} \right\}} \right\rbrack}} \\ {\approx {{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times A} \right)}} \equiv B_{OP}} \end{matrix} & (34) \end{matrix}$

That is to say, ΔR/R satisfying the above expression (34) is the threshold point B_(OP). In this sense, approximation is performed by utilizing constant numbers K_(O) and A that can be handled as a sufficiently great value in the assumed range.

The threshold point B_(RP) occurring when the switch SW_(O) is opened and the switch SW_(R) is conductive in the bias current switching circuit 70 of FIG. 3 will now be discussed.

Here, the threshold current is expressed as the above expression (33) and the sensor bias current I_(BR) is expressed as the following expression (35) with the use of the expressions (23), (24), (27), and (28).

$\begin{matrix} \begin{matrix} {I_{BR} = {I_{TR} - I_{CONT}}} \\ {= {{I_{S}/K_{R}} - {I_{S}/A}}} \end{matrix} & (35) \end{matrix}$

The threshold point B_(RP) at this point is obtainable in the similar manner by replacing the current mirror ratio 1/K_(O), of the sensor bias current generating circuit 50, included in the expressions (30) to (34), with 1/K_(R). ΔR/R≈1/(2×K _(R))−1/(2×A)≡B_(RP)  (36)

The hysteresis width |BH|, which is produced by switching between the switch SW_(O) and the switch SW_(R) in the bias current switching circuit 70, will now be discussed.

At this point, the hysteresis width |BH| is expressed by the following expression (37).

$\begin{matrix} \begin{matrix} {{{BH}} = {{B_{OP} - B_{RP}}}} \\ {= {{\left\{ {{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times A} \right)}} \right\} - \left\{ {{1/\left( {2 \times K_{R}} \right)} - {1/\left( {2 \times A} \right)}} \right\}}}} \\ {= {{{1/\left( {2 \times K_{O}} \right)} - {1/\left( {2 \times K_{R}} \right)}}}} \end{matrix} & (37) \end{matrix}$

FIG. 6 depicts the relationship of the threshold points B_(OP) and B_(RP), the hysteresis width |BH|, and the coefficient 1/A, which are obtained from the above expressions (34), (36), and (37).

What are important in the above expressions (34), (36), and (37) obtained according to the above embodiment and FIG. 6 are that the threshold points B_(OP) and B_(RP) are dependent on the coefficient A, but the hysteresis width |BH| is not dependent on the coefficient A. In addition, since the above expression (28) exhibits that since the coefficient A depends on the resistance ratio, the threshold point is variable by changing at least one of the resistors. The resistor R_(C) is illustrated as a variable resistor in FIG. 3 and FIG. 4, but a resistor R_(A) or a resistor R_(B) may be a variable resistor.

Also, in changing the threshold point, the resistor R_(A) is changed when V_(CC) is set as a reference and the resistor R_(B) or the resistor R_(C) is changed when GND is set as a reference. Accordingly, the threshold point is variable without depending on the reference voltage. In this sense, the expressions (22) and (24) exhibit that the resistor R_(S) cannot be changed because the resistor R_(S) has a coefficient depending on the coefficients K_(O) or K_(R). The above expression (37) also exhibits that the hysteresis width |BH| is not dependent on the sensor drive voltage V_(CC).

The above coefficients K_(O) and K_(R) are given by the resistance ratio. Accordingly, when the coefficients K_(O) and K_(R) are determined, the hysteresis width |BH| is determined to be a single value with no variations in its value, no changes according to temperature, and no changes over time.

Therefore, the sensor threshold circuit according to an aspect of the present invention achieves the hysteresis width that is not dependent on the change in the threshold point.

Third Embodiment Configuration of Third Embodiment

FIG. 7 illustrates a circuit diagram indicative of a configuration of a sensor threshold circuit according to a third embodiment of the present invention.

The sensor threshold circuit includes: a four-terminal sensor 10; a voltage comparator 20; a sensor drive current detecting circuit 30; a sensor drive voltage source 40; a sensor bias current generating circuit 50; a threshold adjusting current generating circuit 60; and a bias current switching circuit 70. The four-terminal sensor 10 is a resistance type four-terminal sensor, and is any one of Hall elements, magnetic resistive elements, strain sensors, pressure sensors, temperature sensors, and acceleration sensors, for example.

In the sensor threshold circuit, a sensor bias current I_(B), which has been generated by the sensor bias current generating circuit 50 and the threshold adjusting current generating circuit 60, is applied to an output terminal of the four-terminal sensor 10, so that a hysteresis characteristic is available at a sensor output voltage V_(SO) (=V_(P)−V_(N)), by use of a voltage drop resulted from a product of I_(B)×R_(OUT), where I_(B) represents a sensor bias current and R_(OUT) represents an impedance of an output terminal V_(P) of the four-terminal sensor 10.

In the sensor threshold circuit with such a configuration, a sensor drive current I_(S) is detected with the use of a resistor R_(S) of the sensor drive current detecting circuit 30, so a threshold current I_(T), which is 1/K (K>0) times of the sensor drive current I_(S), is generated at the sensor bias current generating circuit 50. The sensor bias current generating circuit 50 is composed of: a resistor R_(S); a resistor R_(R) or a resistor R_(O) connected by the control of the bias current switching circuit 70; an operating amplifier 52; and a PMOS transistor 51. In this sense, K has two values due to the control of the bias current switching circuit 70, so the sensor bias current generating circuit 50 generates two threshold currents I_(TO) and I_(TR).

A threshold adjusting current I_(CONT), which is 1/A (A>0) times of the sensor drive current I_(S) is produced at the threshold adjusting current generating circuit 60. The threshold adjusting current generating circuit 60 is composed of: a resistor R_(A); an operating amplifier 61; a PMOS transistor 63; an NMOS transistor 66; and an NMOS transistor 67.

The threshold current I_(T) and the threshold adjusting current I_(CONT) are subtracted at the threshold adjusting current generating circuit 60, so as to generate the sensor bias current I_(B). In the third embodiment, the description will be given of the sensor bias current I_(BO) used as current, in which the threshold adjusting current I_(CONT) is subtracted from the threshold current I_(T). However, the sensor bias current I_(BO) may be current in which the threshold adjusting current I_(CONT) is added to the threshold current I_(T).

Since operations and effects according to the present embodiment of the present invention are same as those of the first and second embodiments, the description thereof will be omitted. Only the above expression (28) is different. The threshold points B_(OP), the threshold point B_(RP), and the hysteresis width |BH| in the sensor threshold circuit of FIG. 7 are considered in the same manner, by replacing 1/A defined in the above expression (28) with the following expression (38). 1/A=R _(S) /R _(A)  (38)

Here, for brief description, it is assumed that the NMOS transistor 66 and the NMOS transistor 67 have the same electrical characteristics.

In addition, the above expression (38) exhibits that the coefficient A depends on the resistance ratio. Accordingly, the threshold point is variable by changing a single resistor.

With the configuration described heretofore, as with the first and second embodiments of the present invention, when the coefficients K_(O) and K_(R) are determined, the hysteresis width |BH| is determined to be a single value with no variations in its value, no changes according to temperature, and no changes over time. Therefore, the sensor threshold circuit according to an aspect of the present invention achieves the hysteresis width that is not dependent on the change in the threshold point. 

1. A sensor threshold circuit that outputs a digital signal with a hysteresis characteristic, in receipt of an input from a sensor, the sensor threshold circuit comprising: a voltage comparator that digitizes an output voltage of the sensor; a sensor drive current detecting circuit that detects a sensor drive current; a sensor bias current generating circuit that generates a threshold current, wherein the magnitude of the threshold current is 1/K (where K>0) times the magnitude of the sensor drive current that has been detected by the sensor drive current detecting circuit, and wherein the threshold current is based upon the output from the voltage comparator; a threshold adjusting current generating circuit that generates a threshold adjusting current wherein the magnitude of the threshold adjusting current is 1/A (where A>0) times the magnitude of the sensor drive current that has been detected by the sensor drive current detecting circuit, wherein the threshold adjusting current generating circuit adds or subtracts the threshold adjusting current to or from the threshold current that has been generated by the sensor bias current generating circuit to generate a sensor bias current, and wherein the threshold adjusting current generating circuit supplies the sensor bias current to an output terminal of the sensor.
 2. The sensor threshold circuit according to claim 1, further comprising a bias current switching circuit that switches between a first resistor and a second resistor, based upon an output from the voltage comparator, wherein the sensor bias current generating circuit generates the threshold current, based upon the first resistor or the second resistor connected by the bias current switching circuit.
 3. The sensor threshold circuit according to claim 2, wherein the threshold adjusting current generating circuit includes: a first resistor for changing a threshold point with a sensor drive voltage used as a reference; second and third resistors for changing the threshold point with GND used as the reference; and an operating amplifier, and said I/A satisfies the equation: 1/A=(R _(S)/(R _(A)/2))×(I−R _(C) /R _(B)) where R_(A) represents a resistance value of the first resistor, R_(B) and R_(C) (where R_(B)>R_(C)) represent resistance values of the second and third resistors, respectively, and R_(S) represents a resistance value of a resistor in the sensor drive current detecting circuit.
 4. The sensor threshold circuit according to claim 3, wherein at least one of the first, second, and third resistors is a variable resistor.
 5. The sensor threshold circuit according to claim 1, wherein the threshold adjusting current generating circuit includes: a first resistor for changing a threshold point with a sensor drive voltage used as a reference; and an operating amplifier, and said 1/A satisfies a following expression, 1/A=R _(S) /R _(A) where R_(A) represents a resistance value of the first resistor, and R_(S) represents a resistance value of a resistor in the sensor drive current detecting circuit.
 6. The sensor threshold circuit according to claim 5, wherein the first resistor is a variable resistor.
 7. The sensor threshold circuit according to any one of claim 1 to claim 6, wherein 1/K=R_(S)/R_(O) is formed when only the first resistor is conductive, and 1/K=R_(S)/R_(R) is formed when only the second resistor is conductive, where R_(O) represents a resistance value of the first resistor and R_(R) represents a resistance value of the second resistor.
 8. The sensor threshold circuit according to any one of claim 1 to claim 6, wherein the sensor is a four-terminal sensor and is any one of a sensor using a Hall element, a sensor using a magnetic resistive element, a strain sensor, a pressure sensor, a temperature sensor, and an acceleration sensor. 